Data transmission method and system

ABSTRACT

Equalizers having variable coefficients which develop correction values from sums of weighted equalizer output errors that enable, among other things, rapid universal convergence both for initial setup and in the presence of data and inexpensive instrumentation. Disclosed are use of timed samples as the basis for the equalizer operation, digital filters as well as tapped delay lines to form the equalizer, weight forming continuously as well as infrequently, weight forming based upon the input to the equalizer, use of weights that have a sliding relationship in which a given error value is weighted by successive weights in forming successive correction values, and error forming based upon subtraction of signals representative of the output desired from the actual output of the equalizer. Equalization in the presence of data is shown using the unequalized input to form the weights. In an example using pseudo-noise the weights are formed by cross-correlation of the input with the pseudo-noise. In an example using no pseudo-noise the data is treated as random and the unequalized input is employed to weight the equalizer output errors directly. Various desired responses are shown helping to solve carrier jitter and other data handling problems.

United States Patent Jerry Lee Holsinger [72] Inventor OTHER REFERENCESwelksleyrMflss- Widrow and Hoff: Adaptive Switching Circuits" 1960 1 1pp 725,312 Wescon Record Part 4 pp. 96- 104 Flkd P 30, 1968 Mantey, P.E.; Convergent Automatic Synthesis f 1971 Procedures for Sampled DataNetworks with Feedback Rel Asslsnee Codex Corporation port SEL 64- 112TR6773- 1 Stanford Electronics f f i Laboratories Stanford, California,October 1964 Continuation-impart of application Ser. No. 573,653 Aug 19,1966, now abandoned. Primary Examiner-Benedict V. Safourek Almmey.lohnNoel Williams 54 DATA TRANSMISSION METHOD AND SYSTEM ABSTRACT:Equalizers having variable coefficients which zc 9 Drawing Figs. developcorrection values from sums of weighted equalizer 52 U S m output errorsthat enable, among other things, rapid universal 1 325/42 convergenceboth for initial setup and in the presence of data m 178/69Ins/65,328,162 333/18 and inexpensive instrumentation. Disclosed are useof timed [51] 3/04 samples as the basis for the equalizer operation,digital filters 3H0 as well as tapped delay lines to form the equalizer,weight Field of Search 178/5, 69, forming continuously as wc asinfrequently weight forming 69 A; 325/38 6333/17 25; 328,162 based uponthe input to the equalizer, use of weights that have 155; 179/2? 235/52a sliding relationship in which a given error value is weighted [56]References Cited by successive weights in forming successive correctionvalues,

and error forming based upon subtraction of signals represen- UMTEDSTATES PATENTS tative of the output desired from the actual output ofthe 3,289,082 11/1966 Shumate 178/66 equalizer. 3,368,168 2/1968 Lucky333/18 Equalization in the presence of data is shown using the 3,375,4733/1968 Lucky 333/ 18 unequalized input to form the weights. In anexample using 3,390,336 6/1968 Di Torro 325/ pseudo noise the weightsare formed by cross-correlation of 3,414,819 12/ 1968 Lucky 328/ 162 Xthe input with the pseudo-noise. In an example using no pseu- 3,4l4,84512/1968 Lucky 333/18 do-noise the data is treated as random and theunequalized ,1 5 2/1969 Lord..... 325/42 input is employed to weight theequalizer output errors 3,440,548 4/1969 Saltzberg 328/155 directly.3,177,349 4/1965 Zaborszky et a1 235/152 Various desired responses areshown helping to solve carri- 3,508,153 4/ 1970 Gerrish et al. 333/18 Xer jitter and other data handling problems.

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SHEET UH 0F 12 q LINE HZ?! PATENTEUncI 19 I97! SHEET OSUF 12 PATENTEDucf19 197i SHEET 070F 12 I'll-l PATENTEDum 19 |97| sum 10 or 12 SHEETl1UF12 NvN PATENTEDUBI 19 ml SHEET 12 [1F 12 ll. u-rl will ll. lll llWWX \WNmy MQ HL! DATA TRANSMISSION METHOD AND SYSTEM This application isa continuation-in-part of applicant's copending U.S. Pat. applicationSer. No. 573,653 filed Aug. 19, 1966, and now abandoned.

This invention relates in general to equalizers and to data transmissionsystems and more particularly to high speed electrical pulse systemwhich has operating characteristics and features that make itparticularly useful for the rapid transmission of data pulses overtelephone lines or undersea cables. The system of this invention mayalso be used for transmitting data pulses, between a transmitter and areceiver, over links other than a telephone line.

Telephone lines and cables provide readily available and reasonablyinexpensive links for the transmission of electrical signals. But theselines have been developed over the years to be suitable for voicetransmissions and their electrical and operating characteristics areextremely poor for the transmission of data. Their range of frequencyresponse is low, in the neighborhood of only several thousand cycles persecond. They cause a phase distortion in the transmitted signal. Theiramplification characteristics are not uniform between one line andanother or on the same line over a period of time. Signals transmittedover telephone lines are frequently multiplexed and otherw'ue processedduring transmission so that the transmitter frequency is not accuratelyreproduced at the receiving end of the line and the error is not uniformor constant with either time or frequency.

The system of the present invention concerns a transmitter and areceiver, which are sometimes referred to in the art as a modem. Theterm "modem" was apparently derived from the terms modulator-demoduhtorbut it is now applied to the transmitter-receiver generally. The modemdescribed herein, that embodies this invention, has been especiallydeveloped to be interconnected by a telephone line. The telephone lineis thus a link between the transmitter and receiver. When speaking ingeneral, the means over which data is transmitted from transmitter toreceivershall be referred to as a link.

Presently known modem systems pose problems in being adapted to thediffering electrical characteristics of these telephone lines. Theyoperate too slowly. The rate at which data pulses can be accuratelytransmitted and received is limited as compared with the speeds desiredfor this type of communication.

It is thus a major purpose of this invention to provide a system whichwill make it possible to increase the rate at which data can betransmitted over circuits and links with poor transmittingcharacteristics and, in particular, to enable telephone and cablecircuits with wide ranges of such electrical characteristics to serve aspractical links for high speed data pulse communication.

It is a very specific goal of this invention to provide a system ormodern which will transmit 9.6 kilobits of information per second overwhat are known as schedule 48 telephone lines.

Because no two telephone lines are alike, because any given telephoneline changes with time (sometimes quite rapidly) and becauseitisimportanttostandardiseontransmittingand receiving equipment, it is afurther object of the present invention to provide a system capable ofadapting itself to achieve high speed data transmission rates with linesor connecting links having different characteristics. Indeed, while thefollowing discussion will for convenience and simplicity refer to theapplication of the system or modem to a telephone line, including thevarious types of telephone transmission connection circuits, it shouldbe understood the invention is not limited to th': particularapplication. The portion of the modem that functions to bring about thisself-adaptation of the system to the link is called an adaptiveequalizer.

Known modems employed with telephone line links adapt to the wide rangeof characteristics found in ordinary telephone lines by employingadjustable all-pass network equalizers in which the phase distortioncharacteristics are manually juggled on ,a trial and error basis by anoperator in order to compensate for the distortion in the link. Thismanual setup of the equalizer requires a great deal of time to obtaineven approximate equalization. As a practical matter even thisapproximate equalization is not realizable because of the time factor.Accordingly, it is another purpose of this invention to provide a systemor modern which adjusts itself to the characteristics of the particularlink with which the modem is used.

It is another purpose of this invention to have such adjustment occurfrequently to compensate for the variations in phase and other linecharacteristics during the transmission of data.

It is another purpose of this invention to provide a technique for theadjustment of the compensating characteristics of the receiver whiledata is being transmitted and without requiring the interruption of datatransmittal. The electrical characteristics of the line-frequentlychange during transmission. It is therefore important that the inventionprovide a frequent or substantially continuous adjustment of thereceiver so as to follow'and adapt to changes in the linecharacteristic.

Other purposes and objects of this invention will become apparent from aconsideration of the following discussion.

As a brief introduction, the nature and general teaching of theinvention can be summarized as follows.

It is well known in the art to provide for the rapid transmission ofelectrical data pulses over expensive connecting links that have goodelectrical characteristics. Over such links there is tolerably littledistortion, and such transmission has many uses, indeed, it takes placewithin as well as between the various parts of most data processingequipment.

The problem arises when the machine parts handling the data are widelyseparated and an attempt is made to use readily available (and thusreasonably inexpensive) lines. Those lines cause distortion in thesignal that, among other things, forces the use of big, clear and widelyseparated information pulses that can be easily detected at the receivereven though they may have been severely distorted. But the use of suchpulses automatically means a slow data rate.

According to an important aspect of the present invention, there areprovided improved ways of detecting the various distortions that mayoccur, of estimating those distortions, and of then adjusting thereceived pulses to compensate for the distortions. An equalizer isprovided that has a set of variable parameters (often referred to ascoeflicients" iftheir disposition in the equations defining theequalizer operation makes that term appropriate) for filtering a signalstream. Error values in the equalizer output are determined andsuccessive corrections are added to the coefficients, the correctionsbeing formed by, for each correction value, establishing a set ofindividual error values, weighting each with a particular weight, andforming the correction value as the sum of the weighted error values.

ln briefest terms, a preferred embodiment of this invention involves theuse of a digital filter to equalize (that is, compensate for distortion)the received pulse train. A known pseudonoise signal is transmitted withthe data signal to provide a reference so that the distortion can bemeasured or at least estimated. After having been passed through thedigital filter, the received pseudo-noise signal is cross-correlatedwith a locally generated pseudo-noise reference signal to provideestimates S, of the errors in the digital filter output. These estimatesS, are weighted by weights H, and combined to provide correction factors(a)C, for each coeflicient X,. The estimated error S, values arecombined so that each S, value has an influence on each correctionfactor (a)C,. Furthermore, the weights H, are developed to reflect theextent of intersymbol interference caused by the distortions in thelink. In calculating each correction factor (a)C,, not only is eachestimated error S, employed, but a comparable number of the H, valuesare also employed. Thus a large number of the weights H,.have aninfluence on each correction factor (a )C,. The exact interrelationbetween these various X,, S,, C, and H, values can only be stated by aset of equations. FlG. 2A shows the set of equations that apply to apreferred embodiment.

The correction factors (a)C, are applied to modify the values of thecoefficients X, in the digital filter and the above process repeatedcontinuously. When equalization has been achieved each estimated error8, will average to zero and each coefficient X, will stabilize aroundits own fixed value.

Another important aspect of this invention comprehends the continuedadjustment of the coeflicients while transmitting data (without the useof a reference signal) by determining error values based upon recovereddata values, taking a function of the error values and weighting bysamples of the equalizer input, and forming the correction values as afunction of the sum of the weighted terms. Included is a setup mode inwhich a random pulse sequence is substituted for the data, to achieverapid initial setup of the coefficients.

Still another aspect of the invention is the adaptation of any of thesetechniques to desired equalizer responses which provide data pulsespectrums which are useful for handling problems such as phase jitter,timing recovery and multiplexrng.

Other objects, purposes and features of this invention will becomeapparent from a consideration of the following detailed description anddrawings in which:

FIG. 1 is a diagrammatic representation of a tapped delay line employedas an equalizer. FIG. 1A illustrates the problem tail that may appearunder certain conditions.

FIG. 2 is a block diagram of a system embodying a preferred form of thisinvention. This system is the modem (transmitter plus receiver) andlink. The receiver includes the adaptive equalizer. The equations ofFIG. 2A indicate the significant functions performed by the circuits ofFIG. 2.

FIG. 3 is a basic block diagram of an adaptive equalizer incorporatingthe teachings of this invention in a preferred form, which preferredform involves the employment of digital filtering. FIG. 3 is asimplified block diagram of the right-hand portion of FIG. 2 and thusthe equations of FIG. 2A apply to FIG. 3 in the manner indicated on FIG.3.

FIGS. 4 and 5 are block diagrams of the transmitter section and thereceiver section, respectively, of an adaptation of the FIG. 2 system tothe transmission of data over telephone lines. These two block diagramssimply show the additional circuitry necessary for converting thetelephone line link into the appropriate low pas filter that is presumedto be available as the link between the transmitter and receiver in FIG.2.

FIG. 6 is a block diagram of a jitter control circuit that is desirableto employ when this invention is applied to a telephone link. FIG. 6A isa frequency distribution diagram of the signals employed in transmittingover a telephone link.

F IG. 7 is a generalized block diagram illustrating the weightingtechniques of the invention.

FIG. 8 is a block diagram, similar in form to FIG. 2, of a systemembodying a further preferred form of the invention.

FIG. 8a is a block diagram of a modification of FIG. 8 to illustrate amore general form of the invention for eq' g in the presence of data.

FIG. 8b is a block diagram similar to FIG. 8, showing a preferred setupimplementation of the equalizer of FIG. 8a.

FIG. 9 is a basic block diagram, similar in form to FIG. 3, of

an adaptive equalizer incorporating the teachings of this invention inanother preferred form. It is a simplified block diagram of the righthand portion of FIG. 8.

FIG. 10 is a block diagram, similar to the right portion of FIG. 8, of afurther preferred form of adaptive equalizer incorporating the teachingsof this invention.

FIG. 11 is a frequency distribution diagram of signals;employed intransmitting over a telephone link in which the desired response of anequalizer and the pilot tones have been related so as to improve theability to recover data.

FIG. 12 is a diagram similar to FIG. 11 in which another preferreddesired response is employed.

FIG. 13 is a circuit diagram similar in form to FIG. 5, showing aninstrumentation of the desired response of FIG. 12 to combat phasejitter.

FIG. 14 is a diagram similar to FIG. in which an example of a preferredclass of desired responses of an equalizer is employed.

FIG. 15 is a general block diagram for an equalizer for the class ofdesired responses illustrated in FIG. 14.

FIG. 16 is a general block diagram of a multiplex system employing theequalizer desired response illustrated in FIG. 12.

The description is intended to be read in light of what is already knownto those skilled in the art, for example, that material described andreferred to in the Handbook of Automation Computation and Control,particularly Volume 2 and the publications referred to therein, editedby Grabbe, Ramo & Wooldridge, published by Wiley (1959); R. W. LuckyAutomatic Equalization for Digital Communication," Bell System TechnicalJournal, Apr. 1965, pp. 547-589; F. K. Becker, et al., AutomaticEqualization for Digital Communication," Proceedings IEEE, Jan. 1965,pp. 96-97; M. A. Rap peport, Automatic Equalization of Data TransmissionFacility Distortion Using Transversal Equalizers," IEEE Trans. Comm.Tech., Sept. I964, pp. 65-73; K. E. Schreiner, Automatic DistortionCorrection for Efficient Pulse Transmission," IBM Journal, Jan. 1965,pp. 20-30; W. S. Mohn, Jr. and L. L. Steckler, Automatic Time-DomainEqualization, Ninth National Comm. Symposium, Utica, N. Y., Oct. 7-9,1963, pp. 1-9; G. K. McAuliffe, ADEM-An Adaptively Data Equal ized HighSpeed Modem," International Conference on Military Electronics, 1964,MIL-E-Con 8, pp. 332-337. As a convenience to the reader, the followingdescription includes some more specific references where it is thoughtthey might be helpful, but such references are limited in number in theinterests of brevity.

FIG. 1

FIG. 1 is a diagrammatic representation of a tapped delay line 100 usedin an equalizer. There are known systems that incorporate the FIG. 1equalizer technique. The purpose of FIG. 1 herein is to provide asimplified explanation of the general technology with which thisinvention is involved.

In FIG. 1, a sequence of amplitude modulated information pulses Z(t) isapplied as the input to a low pass filter 105. The output of the lowpass filter is a distorted series of pulses Y(r). The low pass filter105 (which includes a link between the transmitter and the receiver) hasa transfer function b(f). If the receiver could include a circuit whosetransfer function were the inverse 'of'bq), then the distortions imposedby the low pass filter 105 could be compensated and accurate outputpulses Z(t) provided. It is simply not possible to take such an approachto solving the problem of compensating for the distortion in thetransmitted signal for a number of reasons. The basic reason is that aninverse filter would greatly amplify line noise. Furthennore the cost ofbuilding such an inverse filter would be prohibitive.

The kind of distortions imposed by a typical low pass filter 105 (wheresuch filter may include a telephone line) is suggested in FIG. 1 by acomparison of a single input test pulse 2, with the corresponding outputpulse Y, from the filter 105. The corresponding output pulse Y, is sodistorted that it is not only rounded but it is also preceded andsucceeded by a series of what look like damped oscillations, that areconveniently called tails. When a series of pulses Z(t) are the input,the output Y) is the sum of a series of complex waves spaced from oneanother by the period between pulses in the input series Z(t). Themeasured value of each main output pulse in the train Y(t) isappreciably afiected by the tails of the preceding and succeedingpulses. This intersymbol interference is a major limitation on the rateat which data can be sent.

These pulses 2(1) have a uniform time period of T seconds. Specifically,one pulse 2,,(1) comes along once each T seconds. If T equals l/3200thof a second then the pulse repetition rate is 3.2 kilocycles. it ispreferable, and it will be assumed throughout this specification thatthese information pulses 2(1) are adjacent to one another sozero-amplitude period between pulses.

sample instants is called equalization. it might be worth keepequalizerwhich performs this equalization, is an inverse filter over thefi'equency range zero to onehalf T c.p.s.

of the center tap are designated as X, and X By this notationalconvention, the gains of the two amplifiers at the end taps aredesignated X. and X There are thus, 2n+l taps and 2n+l tap gainamplifiers 110.

The combined output of the amplifiers 110 is a continuous signal P(r).it has been found, and is well known in this art, that the appropriateselection of gain values X, for the various amplifiers l 10 will providean output signal PU) which, ifsampied once each T seconds, at theappropriate time each '1' seconds, will provide a series of outputvoltage magnitudes P sampling function. it is closed for an instant onceeach T seconds and thus provides an output pulse P, where t=kT.

The tapped delay line 100 and series of tap gain amplifiers described inLandee, Davis, Albrecht, Electronic Designers Handbook, McGraw-Hill,Inc., 1957, pp. -59 to 20-61.

The output of each tap is supplied to a variable gain amplifier "0. Thefunction of these amplifiers 110 is to vary the amwhich they areconnected. They In practice and for economy, the ordinary variable gainamplifiers can be replaced by analog-to-digital converters, digitalpotentiometers, and inverting amplifiers which can be used to serve thesame function. For a discussion of such potentiometers, see Grabbe, Ramoand Wooldridge, Volume 2, supra, at pp. 20-50 to 20-60.

The tap outputs, afier'each has been adjusted by its variable gainamplifier 110, are summed to provide the signal P(r) that is sampled bythe switch 115 and provide a sample output P, once every T seconds. a

The gain X, of each variable gain amplifier 110 is referred to herein asa tap gain. it has been known to determine the tap gain X, settings byapplying an iterative rule to a sequence of widely spaced test pulses2,. One example of an iterative rule that can provide satisfactory gainX, values is:

wherein: Xf= tap gain at i th tap X '==corrected tap gain atith tap a ascale factor having a magnitude less than 0.1

1', value of i th sample when the gain at that X,= tap gain at thecenter tap that there is no P, value of the center sample 2, the knownvalue of the test pulse.

The above rule was known prior to this invention and is applied to FIG.1 herein in order to set the background for an applied to analog gestedin FIG. 1.).

A typically nearly equalized signal P, representing a single pulse 2, isshown in FIG. 1 together with an indication of the value P to P, thatare supplied as sample outputs P In this desired response, each P exceptfor P,, should be zero, and if the tap gains X, are properly selectedsuch a state can be achieved. When the output is such that each P, iszero and'P, equals Z, then the receiver is said to have been equalized.The value of each sample P, is influenced by each tap sample is afiectedby each dominant effect on the sample P The sample P itself, is anoutput error since ideally, and when equalized, P is zero. However, thetap gain X error is not proportional to the P value, because the P valueis influenced by all the tap outknown rule mentioned above, ascorrection for the error of the corresponding tap gain X If a smallcorrection (a) P proportional to the magnitude of that error P is madeto the tap gain X and a comparable operation performed on each tap gainX,, then a new set of tap gain Xf values are obtained. The reference tapgain, which inthiscaseisthecentertapgainhhastobeadjustedbased on thedifi'erence between the sample P, value and the known value of the testpulse 2,. Thus the new set of tap gains Xf" bear the relationship to theold set of tap gains X, that is expressed in the above equation, namely:

This process is repeated. Another test pulse 2, is sent. equalized andsampled. The new sample values are used for another set of tap gain X,corrections (a)P,. The process is continued until the tap gain X, valuesconverge to final values, at which point full equalization is achievedand the sample P, is equal to the test pulse Z, value.

if data pulses Z(t) are now sent, they will be properly equalized andthe output sample values P will accurately represent the data pulse 2(1)values.

It should be noted however that there are conditions using the abovementioned known rule under which the X, values will not converge;primarily when the phase distortion imposed by the low pass filter isvery severe. it is one of the purposes of this invention to provide amechanism and method for assuring convergence no matter how bad thephase characteristics of theline may be.

With the above basic concept of the problem involved in mind, it will beeasier to comprehend the following discussion of FIGS. 2 and 3, whichfigures illustrate an embodiment of this invention.

problems with this approach.

First, a constant data flow to the transmitter and from the receiver isdesired. Since telephone line characteristics change rapidly enough torequire readjustment of the tapped delay line every few minutes, thisimplies that data buffering is required. Although this is not aninsurmountable problem, the complexity of such bufi'ering is quitesignificant especially when suitable measurement averaging isincorporated to reduce the effects of telephone line noise. Moreover, itis usually necessary that a feedback channel be used to instruct thetransmitter when to interrupt data and send the reference signals P,.

Second, in using this technique, the telephone line sees a high dutycycle signal when data is transmitted (i.e., a signal with a lowpeak-to-average power ratio) while it sees a very low duty cycle signalduring the equalization time. This is undesirable since slight linenonlinearities caused by AGC circuits, compandors, modulators, and otherpotentially nonlinear components can cause errors in the resulting tapgain X; settings.

FIG. 2-General The basic system or modem that incorporates thisinvention is shown in FIG. 2. FIG. 2A is a mathematical description ofthe operations shown in block diagram form in FIG. 2. The FIG. 2 modemwhich accomplishes the equalization discussed above employs variousdigital techniques toachieve equalization. A digital filter is employedto equalize the received information signal. The settings of the digitalfilter are adjusted and adapted to the link over which the informationis being sent by various digital techniques. A reference signal (thepseudonoise) is digitally cross-correlated to provide estimates of theerrors in the digital filter. Corrections to the digital filter are madeby processing these estimated errors through digital circuitry thatweights and recombines the estimated errors to provide correctionfactors. The equalization thereby achieved is analogous to that which isachieved by use of a tapped delay line.

The FIG. 2 embodiment is described discussion of the transmitting andreceiving unrts up to the point where the digital adaptive equalizingtakes place and then by a more detailed description of the adaptiveequalization and the units employed to provide equalization and selfadaptation of the equalizing to the link.

The low pass filter 200 includes the link between transmitter andreceiver. As will be described further on, this link may be a telephonesystem. The telephone system may be converted to a low pass filter bymeans which are well known in this art and are described in connectionwith FIGS. 4 and 5. Suffice it at this point to accept the fact that alow pass filter 200 is the link between transmitter and receiver.

One of the notational conventions employed in connectionwith the figureshas to do with the manner in which various frequencies and time periodsare shown. A master oscillator 202 at the transmitter and another masteroscillator 204 at the receiver are both employed to provide a masterfrequency signal, in the embodiment shown, of 2.4576 megacycles. Thismaster frequency signal is applied to various counter and timingcircuits 206 and 208 in order to provide the various signals necessaryto operate the units shown in the block diagram. In the embodimentshown, the information pulses (data and pseudo-noise) are generated at a3200 pulse per second rate. Thus many of the operations that have to beperformed require a 3.2 kc. signal 1/3200 seconds. The and timingcircuits 206 and 208 provide this basic frequency. This basic frequencyis indicated in the figures asf, and this fundamental time period of1/3200 seconds is indicated as (T). For example, the sampler 252 istriggered by an input signal f, in order to provide a sample Y., of thecontinuous input signal Y(t) once each T seconds. The

output from the sample 252 is designated as Y. and the periodicity ofoutput by (T). A corresponding notation is used throughout in which theoutputs and inputs for most of the blocks shown are designated by anotation as to their meaning as well as a notation of the periodicityfor each pulse or value. The triggering frequencies for most of theblocks are also shown and, it will be noted, that the triggeringfrequencies are by and large multiples of the basic 3.2 kc. data ratefrequency. Indeed, the master oscillator outputs of 2.4576 me. are amultiple of the basic data rate frequency of 3.2 kilocycles.

It should be understood that in the operation of the various digitalcircuits other signal frequencies will be required. For example, certainsignals required will be a function of the number of bits employed incarrying each of the values in any particular multiplier or memory unitSuch matters as this are within the knowledge of those skilled in theart and will not be dealt with in any detail herein.

At the transmitter, the master oscillator 202 establishes the basicfrequency from which the various frequencies necessary for the operationof the system are derived. in the embodiments shown herein the basicfrequency is 2.4576 megacycles per second. The master oscillator 202drives a timer counter 206 which produces the required synchronizationand pilot frequencies at the proper phase relation to each other.

Data in binary form is supplied for transmission at the rate of 9600bits per second from a data source 210. A digital to analog converter212 converts this data to an eight-level data pulse sequence, therebybringing the pulse rate down to 3.2 kc. This means that each three databits are encoded onto a data pulse, the height of the data pulse beingthe encoding means. There are eight posible combinations of three bitsand thus an eight-level data pulse can be used to encode three bits. Apseudo-noise generator 214 provides a basic reference pulse signal. Thisreference train of pulses of known, predetermined, sequence and heightare the reference against which the receiver can calculate the adaptiveadjustments needed to provide an accurate output. By adapting thereceiver to provide an accurate reference pulse train output, it followsthat the data pulse train output will also be accurate.

The reference pulse signal sequence is in the general fonn of atwo-level sequence and. in particular, is one of the pseudo-noisesequences given in Appendix 11, Table 2, p. 169, of Golomb, Baumert,Easterlirng; Stifl'ler, and Viterbi, Digital Communications with SpaceApplications," Prentice-Hall, Englewood Clifi's, N.J., 1964, Chapters 1and 8 and Appendix I]. This sequence may be generated as described inthese pages and in view of what is already known to the art. While forthis embodiment the sequence that is 63 pulses long is preferably used,it is believed that any two-level sequence as defined at pp. 51-52 ofthis work may be used.

Thus the generator 214 provides a 3.2 kc. pseudo-noise pulse train,which train is repeated each 63T seconds. The amplitude of thesepseudo-noise pulses conveniently may be :t1 [2 (i.e., 00.10 or 1 1.10 inthe standard twos complement form of binary notation). The two levelpseudo-noise pulses (one each T seconds) are added to the a composite3200-pulse transmitted through the eight level data pulses to provideper second infonnation signal that is low pass filter link 200. The lowpass filter link 200 distorts this information signal to provide adistorted information signal Y(t) as the input to the receiver.

A sinusoidal 1.6 kc. pilot tone, employed for synchronization purposes,is also transmitted with the information pulses. This pilot tonecontrols the frequency and phasing of the master oscillator 204 at thereceiver by means of a known phase locked loop type of operation. Themaster oscillator 204 is a voltage controlled oscillator having a 2.4576megacycle center frequency. A 1536:] counter 216 provides a 1.6kilocycle output when the master oscillator 204 is properly putting outits 2,4576 mc. signal. The two 1.6 kc. pilot tone and 1.6 kc. counter216 output signals are the two inputs to the phase discriminator 218.The phase discriminator 218 dutput thereby locks the master oscillator204 (and thus all the counter and timing circuit 208 outputs) to thephase of the master oscillator 202 at the receiver.

The sampler 252 in this embodiment is timed as follows. In particular,it is not operated to sample at the peak amplitude of the receivedpulses Y(t). Instead, the 1.6 kc. pilot tone frequency output from thecounter 216, which has been phase shifted 90 for the purpose of beingsupplied to the phase dis criminator 218, is phase shifted back at 220to be in phase with that of the received 1.6 kc. pilot tone. it is thenapplied to a zero crossing detector 222 which senses the points at whichthis 1.6 kc. signal (and thus the in-phase pilot tone) goes through zeroamplitude. At these points a triggering pulse is released to the sampler252 to cause it to sample.

The first, and a minor, advantage of this technique for controlling thesampling is that the sampler 252 samples when the pilot tone is zero andthis eliminates the need for a filter. It also allows the pilot to beplaced lower in the spectrum which might be more convenient in someapplications. It also allows width (and below the frequency limit of theconnecting link). This advantage is of particular importance when theinvention is applied to telephone line links as in the FIG. 4 andembodiment.

The more significant advantage of this method of sampling is that itsolves a problem which has plagued the prior art. The problem is thatfor some low pass filters containing as a part thereof a phone line, theequalizers are previously used have succeeded in compensating more orless for the data pulse distortion at the tap outputs but they have leftand in a sense generated a large problem tail. See Lucky, supra, FIG. 9.

The nature of the problem can best be seen by reverting to FIG. 1wherein widely separated reference pulses are presumed to be used (thatis, pseudomoise is not used). The problem tail appears beyond thesamples-that are collected for equalization. Then, when the system isthought to be equalized and data is again transmitted, the tail appearsin the data to cause error. This illustrated in FIG. 1A, in which theP,s are sampled outputs of a test pulse from the equalizer.

This problem is solved in an intuitive way by an operator when the tapgains or gain factors are adjusted manually. The operator observes theoutput of the test pulses, widely spaced, on a scope and, ifthe problemtail is present, then he adjusts the gains by trial and error until,while only approximate equalization is obtained throughout the length ofthe tapped delay line, the problem tail is minimized. This approach isnot suitable in application to an automatic system for, first, theautomatic system attempts to achieve perfect equalization. Second, itwould be cumbersome to automate the procedure of sampling the tail of atest pulse outside the length of the line and adjusting the coefficientsX, so as to strike a good compromise. Furthermore, where the data is notinterrupted in an embodiment using pseudo-noise and adjustments are madewhile data is being received, the tail of a particular pulse in asequence appears in the next repetition of the sequence, along with thenext repetition of that particular pulse in the next sequence and it hasbeen found that, as a general rule, the pseudo-noise equalization cannotequalize both large pulses at once.

It has been thought that the problem has been caused by the samplingtechnique of the prior art which has been to sample at the maximum ofthe peak pulse generated at the receiver by the transmitted pulse. Tosample at the maximum of this peak pulse sometimes is, depending uponthe phase distortion of the low pass filter in question (particularlythe distortion in the vicinity of the frequen yfrn) to sample so as torequire that the digital equalizer (which is in a sense an inverse lowpass filter) have nearly infinite gain at the frequency f,/2). This isphysically impossible and it has been found that the problem tail is aresult The automatic systems of the present invention solve this problemby selecting sampling times by which the tail problem is avoided for alllines in a systematic fashion. These systems operate as follows:

The 1.6 kc. pilot tone is phase controlled so that its zero crossoverpoints at the transmitter are in the center of the data equalizer outputsample 2, values. When pulses. in other words the pilot is phase shifiedabout a quarter cycle of from the data at the transmitter and if thedata has a periodicity of T, the pilot must have a frequency one-halfthat of the data and thus a periodicity of 2T. This frequency must beexactly f /2if it is to be transmitted continuously and used to directlycontrol the timing of the samples. It need only be approximatelyone-half of f, if it is to be used to approximate the phase distortionof a f,l2 frequency caused by the connecting link and if thisinformation is to be indirectly used to control the phase of thesamplers sampling.

Thus, the properly phased sampler 252, samples the continuous incomingsignal Y(r) once each T seconds to provide a voltage magnitude Y(t=kT)once each T seconds. The sampling for all practical purposes isinstantaneous. Instantaneous sampling can be achieved by feeding thecontinuous input Y(t) through a closed switch to a capacitor. Thetriggering signal from the zero crossing detector 222 then opens theswitch at the appropriate instant and the voltage value Y( r=kT isavailable from the capacitor.

The analogue to digital converter 254 then converts this sample valueY(t=kT) into a binary indication Y, of the signal Y(t) magnitude at themoment of sampling.

FIG. 2 Adaptive Equalization The following description of the adaptiveequalization is presented in more detailed form than is the rest of thedescription herein because the rest of the equipment shown or describedis known in the art or, at the least, previously known to others as wellas myself.

The input to the low pass filter 200 includes a 3.2 kc. data signal, a3.2 kc. pseudo-noise signaland a 1.6 kc. pilot tone signal to provide alow pass filter 200 output Y(t). For the purposes of this discussion ofequalization, we can ignore the 1.6 kc. pilot tone. As described above,sampling takes place when the pilot tone passes through zero so that thesample 252 output Y,, is afl'ected only be the data and pseudo-noisepulses. lf sampling is to take place at some other point in the pilottone cycle it is necessary to subtract oil the known errors introducedthereby.

Thus it is appropriate to say that the continuous information signalY(r) is composed of a series of pseudo-noise pulses added to the seriesof data pulses. This infonnation signal ((1) however is distorted byhaving been passed through the link 200. The distorted informationsignal Y(r) is sampled once each pulse period T to provide a series ofsampler 252 outputs Y(t=kT). An analog to digital converter 254 convertsthese l(r=7') information signals to digital values Y, which arereceived and stored in the Y, recirculating memory unit 256.

By a means that is known as digital filtering the sample values in therecirculating memory unit 256 are multiplied by various coeficients X,that are stored in the recirculating memory unit 258. The digitalmultiplier 260 and accumulator 262 perform this digital filteringfunction by multiplying sample Y, values by coefficients X, values andsumming them in the fashion called for by Equation (1) of HG. 2A toprovide the coefficient values X, are correctly derived, the equalizerbe properly equalized and will correspond to the heights of theinformation pulses Z(r) that are produced at the transmitter end of thesystem. The equalization compensates for the distortion caused by thelink 200.

These equalizer output 2,, values contain, in addition to the desireddata, a value corresponding to the level of the pseudonoise signalapplied at the transmitter. Accordingly, a pseudonoise generator 264,which has been synchronized with the transmitter pseudo-noise generator214, is employed to-subtract 06 the pseudo-noise values so as to providedata output.

The equalizer output Z. values are held in a recirculating memory unit266 and, because these 2,, values also include pseudo-noise information,they form the basis for checking out whether or not the coefficients X,have the desired values. This is done by crom-correlating a sequence of2,, values with values are provided.

These estimated errors S, are then weighted by certain preset weights H,according to the procedure called for in and accumulator 274,respectively.

As a consequence, the correction values C, are stored in a C, memoryunit 276. A reduction factor a,", of the order of one -second, isapplied within the C, memory unit 276 to provide appropriately reducedoutput values a(C,) which are the actual correction factors applied tothe corresponding coeflicients X, in the memory unit 258, see Equation(2).

coefficients X, to converge to those values that will provide fullyequalized output 2,.

Important to this convergence is the manner in which the individualweights H, are obtained and this is described in greater detail furtheron.

The X, memory unit 258 has capacity for 29 X, values and thus retains 29coetficients X,.

The Y, recirculating memory unit 256 is designed to circulate 29successive Y, values. By analogy, it is like a tapped delay line having29 taps. Once each '1 seconds, a new Y, value is added and the oldest Y;dropped. Thus each T seconds, each sample Y, value moves up one in thememory unit 256. It also follows that a given Y, input is in the memoryunit 256 for 291 seconds.

The digital multiplier 260 operates so that once each T periods each Tseconds.

The subscript notation in Equation (1) indicates which X, value ismultiplied by which Y, value. For example, the center X, value, i.e., X,(which is the largest coeflicient) always multiplies the center Y,value. Thus a given 2, has as its primary component the Y, X, product,where Y, corresponds to 2 and I4 succeeding Y, values (i.e., Y,,,). inconnection with Equation (1) it should be kept in mind that the 2,,provided at the output of the accumulator 262 at any one instantcorresponds to the Y, that If! seconds earlier was at the Y, position inthe Y, recirculating memory unit 256.

The accumulator 262 stores the 29 X, Y, products generated each Tseconds and adds them together to provide a modified sampled pulseoutput Z, value once each T seconds.

If the coeificients X, in the memory unit 258 are properly set, then the2,, value at the accumulator 262 output will accurately represent theamplitude of the corresponding informa tion pulse that was fed into thelow pass filter 200. That is, equalization will be obtain Since theinformation pulse was m dified in amplitude by one of the pseudo-noisepulses supplied by the generator 214, that modification must be undonein order to provide accurate data output. The pseudo-noise generator 264and subtract circuit 280 perform this function. Subtract circuits ofthis sort 12 are discussed by Grabbe, Ran; and Wooldridge, supra, at pp.8-1 1.

The 29 slots for the memory units 256 and 258 are selected to provideadequate equalization for the range of distortion using larger capacitymemory units 256, 258 and more coefiicients X,.

In Equation (1), the indication X,,,=8,, means that at the outset, thecoeflicients X, are preset so that X,=l and all the rest of the X,values are zero. This condition is required for proper calculation ofthe weights H, as will be described further on.

emu-correlation with the pseudo-noise R, sequence. The 2, output valuechanges once each T seconds.

When the connection between the receiver and transmitter is first made,the coeflicients X, in the memory unit 258 will normally be very muchdifierent than is desired. F urtherrnore, during the course oftransmission, the characteristics of the low pass filter 200 may vary,as is typically the case where telephone lines form all or part of thelow pas s filter 200. In order to arrive at correct X, values and tokeep such X, values continuously revised, the following adaptive processtakes place.

If the coefficients X, were all correct for the condition of the link200, then no further correction would be needed. But as the linkcondition changes, the equalization must be revised.

These estimated errors S, are obtained by the cross-correlation of asequence of output values 2, and the pseudo-noise sequence R accordingto the relationship shown in Equation (3).

Because the pseudo-noise sequence is a repeated 63 pulse sequence, itbecomes necessary to run the cross-correlation that the link 200 noiseand the data (which data is noise to the known pseudo-noise sequence)interference are effectively eliminated. The summation range in Equation(3) is over 16? time periods T, that is 1008 T seconds. The summationrange where It goes from MPV+1 to MP(V+l) is 1008, where P==63 (the timeperiod of the pseudo-noise sequence) and M=l 6.

The value of M need not be 16. But a value of M near 16 appears to be agood compromise between the undesirable con sequences of a much smalleror a much larger M. In either case, that is if M is made either muchlarger or much smaller than its optimum value, the net result is alonger time to obtain convergence of the coefiicients X,. Roughly, thereasons why this happens are as follows.

If M is made very small, then each estimate of S, is poorer and thescale factor must be reduced. Thus many more estimaof errors S, can bemade. Thus many correction cycles means a lot of down-time" calculatingC, values and, in all, a longer time before equalization is attained.

If M is made very large, then we find that even though the requireslarger units and adds cost without speeding equalization.

' 13 In connection with the magnitude of M in Equation (3), two otherpoints should be kept in mind. First, the magnitude of must be keptroughly proportional to the estimate and the more accurate are thecorrection values; thus the larger may the scale factor a be. Second,the optimum value for M is a function of the transmitting power used forpseudo-noise transmission. The M of 16 illustrated was found useful inan embodiment wherein the pseudo-noise power equalled the data power.The greater the pseudo-noise power, the smaller need M be to obtainequivalently accurate esti- Equation (3) shows that the calculation ofestimated error 8, values involves three it might be possi- L valuesgoing to bias by eliminating the data component of the information pulseZ. values.

F urthennore, there is an additional inherent bias term caused by thefact that the 63 because it is an automatically Without animplementation of the second term (the cross-correlation with m it wouldbe necessary to put in a bias term to cancel the inherent pseudo-noisebias.

The R output from the fast pseudo-noise generator must be synchronizedwith the pseudo-noise output from the generator 214 in the transmitter.The manner of synchronization is discussed in connection with theobtaining of the weight H, values since the synchronization must takeplace in order to calculate these H, values.

The method of calculating correction factors (C,) for the 29coefficients X, involves first making 29 error estimates 8,, whichestimates are then weighted in the manner called for by Equation (4) toprovide the correction values 0,.

It is Equation (3), the terms of which are discussed above, thatdescribes the technique employed in FIGS. 2 and 3 to obtain theseestimates errors S,. 75

It is diflicult to give a word description of exactly what it is thatEquation 3) designates. Roughly, Equation 3) indicates that the value ofeach data pulse 2., is used as the basis for multiplying a 29-pulsesequence portion of the 63 pulse pseudo-noise sequence R, to provide 29LR products in each T seconds. Each of the 29 2,3,,, products are addedinto a separate one of the 29 slots in the S, memory 270. Then the29-pulse sequence of R, values which overlaps by 28 R, values theimmediately preceding 29 R,, values. Each T seconds the 29 R. set shiftsby one pulse within the overall 63-pulse sequence. After 63T seconds,the first 29 R, set is repeated.

More particularly, assume that we are calculating the estimated error SIn order to calculate S a series of 1008 data pulse 2,, values aremultiplied one 1008 successive pseudo-noise pulse values R Theparticular pseudo-noise pulse R employed for each multiplication isrelated to the data pulse 2,, by the subscript notation indicated.Assume that in the sequence of 1008 data pulses 2,, under considerationthat the Z, is being multiplied to provide one of the values which goesinto the summation that provides the estimated error S The pseudo-notepulse R employed will be that pseudo-noise pulse added to R at thetransmitter. The number 61 comes from the k-j subscript, that is minus14. In order to make sure that the noise pulse is multiplied against thereceived data pulse 2,, the pseudo-noise generator 268 in the receiverhas to be synchronized to the pseudo-noise generator 214 in thetransmitter.

Since there are 29 estimated errors 8, to be calculated, it is necessarythat at least 29 of the pseudo-noise pulses in the 63- pulse sequence beavailable each '1 seconds for multiplication against each data pulse 2so as to provide one of the termsfor each of the 29 summations thatEquation (3) indicates is required in order to obtain the 29 estimatederrors 8,. A fast pseudo-noise generator 268 is therefore required. Thisfast pseudo-noise generator 268 generates pseudo-noise pulses having aperiod of 1732 seconds. For 29 of these T/32 seconds, the fastpseudo-noise generator 268 generates 29 pseudo-note pulses in sequenceand for the next 3T/32 seconds, the fast pseudo-noise generator stepsahead by the rest of the sequence so as to be able to start all overagain with the nextdata pulse 2 Actually, the fast pseudo-noisegenerator 268 steps ahead by one less or one more than the rest of the63-pulse pseudo-noise sequence since it is essential that the portion ofthe cessive pseudo-noise pulse. in the practical embodiment illustrated,it makes instrumentation easier to run the pseudonoise sequence backwardand to therefore step ahead one less, rather than one more which,although significant to simplify circuitry, is a matter of choice as faras the basic concept is concerned. number of successive A values havebeen operated upon in this fashion, the 29 separate summations provide29 separate estimated errors S There is nothing critical about employing1008 successive 2,, values as illustrated in this embodiment but it isimportant that a number of successive Z, pulses be a multiple of thenumber of pulses in the pseudo-noise sequence.

It might be noted that the H6. 2 block diagram shows the 2,, values andthe R; values as applied directly to the S, recirculating memory unit270 while Equation (3) indicates that multip 'cation and summationfunctions are perfonned in calpowers of two so that, in binary form, allthat is involvetLis shifting the binary point of each 2,, value and thenmultiplying by a +1 or 1 depending on whether R; is positive ornegative. No accumulator is necessary because these 2,, k products arefed directly to the appropriate slots in the S, memory unit 270.

1. In an adaptive equalizer wherein a digital filter having N variablecoefficients Xi is employed to provide equalization and wherein saidfilter is adapted to receive information pulses which are the sum of asequence of data pulses and a sequence of pseudo-noise pulses, saidfilter including means to crosscorrelate the received pseudo-noise pulsesequence after digital filtering with a locally generated pseudo-noisepulse sequence to provide a sequence of estimated error Sj values, theimprovement comprising: A. storage means for storing a set ofpredetermined weight Hi values, B. error weighting means including amultiplier to multiply each estimated error Sj value by a separateweight value to provide a set of N weighted error values. C. andcorrection means connected to correct each coefficient Xi by addition ofa correction Ci, D. said error weighting means and said correction meansconstructed to cooperate to apply said corrections in accordance withthe equations where Sj are the estimates of the errors formed fromequalizer output samples by cross-correlation with pseudo-noise; Hi jare the predetermined weights; i-j N means that the difference between iand j, ignoring the sign (+or-) of the difference, cannot exceed N. 2.The adaptive equalizer improvement of claim 1 further characterized by:cross-correlation means to cross-correlate the unequalized receivedpseudo-noise pulse sequence with a locally generated pseudo-noise pulsesequence to provide a set of at least N of said weight Hi values.